Adaptive Current Control Timing and Responsive Current Control for Interfacing with a Dimmer

ABSTRACT

In at least one embodiment, an electronic system adapts current control timing for half line cycle of a phase-cut input voltage and responsively controls a dimmer current in a power converter system. The adaptive current control time and responsive current control provides, for example, interfacing with a dimmer. The electronic system and method include a dimmer, a switching power converter, and a controller to control the switching power converter and controls a dimmer current. In at least one embodiment, the controller determines a predicted time period from a zero crossing until a leading edge of a phase-cut input voltage and then responsively controls the dimmer current to, for example, reduce current and voltage perturbations (referred to as “ringing”), improve efficiency, and reduce an average amount of power handled by various circuit components.

CROSS-REFERENCE TO RELATED APPLICATION

This application claims the benefit under 35 U.S.C. §119(e) and 37 C.F.R. §1.78 of U.S. Provisional Application No. 61/570,554, filed Dec. 14. 2011, and is incorporated by reference in its entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates in general to the field of electronics, and more specifically to a method and system for providing adaptive current control tinting and responsive current control for interfacing with a dimmer.

2. Description of the Related Art

Electronic systems utilize dimmers to modify output power delivered to a load. For example, in a lighting system, dimmers provide an input signal to a lighting system, and the load includes one or more light sources such as one or more light emitting diodes (LEDs) or one or more fluorescent light sources. Dimmers can also be used to modify power delivered to other types of loads, such as one or more motors or one or more portable power sources. The input signal represents a dimming level that causes the lighting system to adjust power delivered to a lamp, and, thus, depending on the dimming level, increase or decrease the brightness of the lamp. Many different types of dimmers exist. In general, dimmers use a digital or analog coded dimming signal that indicates a desired dimming level. For example, some analog based dimmers utilize a triode for alternating current (“triac”) device to modulate a phase angle of each cycle of an alternating current (“AC”) supply voltage. “Modulating the phase angle” of the supply voltage is also commonly referred to as “chopping” or “phase cutting” the supply voltage. Phase cutting the supply voltage causes the voltage supplied to a lighting system to rapidly turn “ON” and “OFF” thereby controlling the average power delivered to the lighting system.

FIG. 1 depicts a lighting system 100 that includes a leading edge, triac-based dimmer 102. FIG. 2 depicts ideal, exemplary voltage graphs 200 associated with the lighting system 100. Referring to FIGS. 1 and 2, the lighting system 100 receives an AC supply voltage V_(SUPPLY) from voltage supply 104. The supply voltage V_(SUPPLY), indicated by voltage waveform 202, is, for example, a. nominally 60 Hz/110 V line voltage in the United States of America or a nominally 50 Hz/220 V line voltage in Europe. A leading edge dimmer phase cuts leading edges, such as leading edges 204 and 206, of each half cycle of supply voltage V_(SUPPLY). Since each half cycle of supply voltage V_(SUPPLY) is 180 degrees of the supply voltage V_(SUPPLY), a leading edge dimmer phase cuts the supply voltage V_(SUPPLY) at an angle greater than 0 degrees and less than 180 degrees. Generally, the voltage phase cutting range of a leading edge dimmer 102 is approximately 10 degrees to 170 degrees. The leading edge dimmer 102 can be any type of leading edge dimmer such as a triac-based leading edge dimmer available from Lutron Electronics, Inc. of Coopersberg, Pa. (“Lutron”). A triac-based leading edge dimmer is described in the Background section of U.S. patent application Ser. No. 12/858,164, entitled Dimmer Output Emulation, filed on Aug. 17, 2010, and inventor John L. Melanson.

Triac 106 acts as voltage-driven switch, and a gate terminal 108 of triac 106 controls current flow between the first terminal 110 and the second terminal 112. A gate voltage V_(G) on the gate terminal 108 will cause the triac 106 to turn ON and current i_(DIM) when the gate voltage V_(G) reaches a firing threshold voltage value V_(F) and a voltage potential exists across the first and second terminals 110 and 112. The dimmer output voltage V_(φ) _(—) _(DIM) is zero volts from the beginning of each of half cycles 202 and 204 at respective times t₀ and t₂ until the gate voltage V_(G) reaches the firing threshold voltage value V_(F). Dimmer output voltage V_(φ) _(—) _(DIM) represents the output voltage of dimmer 102. During timer period T_(OFF), the dimmer 102 chops the supply voltage V_(SUPPLY) so that the dimmer output voltage V_(φ) _(—) _(DIM) remains at zero volts during time period T_(OFF) At time t₁, the gate voltage V_(G) reaches the firing threshold value V_(F), and triac 106 begins conducting. Once triac 106 turns ON, the dimmer voltage V_(φ) _(—) _(DIM) tracks the supply voltage V_(SUPPLY) during time period T_(ON). Once triac 106 turns ON, triac 106 continues to conduct current i_(DIM) regardless of the value of the gate voltage V_(G) as long as the current i_(DIM) remains above a holding current value HC. The holding current value HC is a function of the physical characteristics of the triac 106. Once the current i_(DIM) drops below the holding current value HC, i.e. i_(DIM)<HC, triac 106 turns OFF, i.e. stops conducting, until the gate voltage V_(G) again reaches the firing threshold value V_(F). The holding current value HC is generally low enough so that, ideally, the current i_(DIM) drops below the holding current value HC when the supply voltage V_(SUPPLY) is approximately zero volts near the end of the half cycle 202 at time t₂.

The variable resistor 114 in series with the parallel connected resistor 116 and capacitor 118 form a timing circuit 115 to control the time t₁ at which the gate voltage V_(G) reaches the firing threshold value V_(F). Increasing the resistance of variable resistor 114 increases the time T_(OFF), and decreasing the resistance of variable resistor 114 decreases the time T_(OFF). The resistance value of the variable resistor 114 effectively sets a dimming value for lamp 12.2. Diac 119 provides current flow into the gate terminal 108 of triac 106. The dimmer 102 also includes an inductor choke 120 to smooth the dimmer output voltage V_(φ) _(—) _(DIM). Triac-based dimmer 102 also includes a capacitor 121 connected across triac 106 and inductor 120 to reduce electro-magnetic interference.

Ideally, modulating the phase angle of the dimmer output voltage V_(φ) _(—) _(DIM) effectively turns the lamp 122 OFF during time period T_(OFF) and ON during time period T_(ON) for each half cycle of the supply voltage V_(SUPPLY). Thus, ideally, the dimmer 102 effectively controls the average energy supplied to the lamp 122 in accordance with the dimmer output voltage V_(φ) _(—) _(DIM).

The lighting system 100 includes a power converter 123 with a resistor, inductor, capacitor (RLC) network 124 to convert the dimmer voltage V_(φ) _(—) _(DIM) to an approximately constant voltage and, thus, provide an approximately constant current i_(OUT) to the constant current lamp 122 for a given dimmer phase angle. The triac-based dimmer 102 adequately functions in many circumstances. The triac-based :dimmer 102 utilizes a “glue” current during the time T_(OFF) to properly charge the timing circuitry. Additionally, electronic dimmers that include controllers, e.g. “smart” dimmers, utilize current during the time T_(OFF) to provide power to the electronic dimmer. Providing the glue current to the dimmer for the time T_(OFF) has conventionally been considered an unavoidable occurrence.

SUMMARY OF THE INVENTION

In one embodiment of the present invention, a method includes predicting a time period during a cycle of a phase-cut input voltage to a power converter system that is expected to occur in advance of a leading edge of the phase-cut input voltage. The method further includes during the cycle of the phase-cut input voltage, actively controlling a decreasing transition rate of a current conducted through a dimmer at least by the predicted time period that is expected to occur in advance of the leading edge of the phase-cut input voltage.

In another embodiment of the present invention, an apparatus includes a controller configured to predict a time period during a cycle of a phase-cut input voltage to a power converter system that is expected to occur in advance of a leading edge of the phase-cut input voltage. The controller is further configured to, during the cycle of the phase-cut input voltage, actively control a decreasing transition rate of a current conducted through a dimmer at least by the predicted time period that is expected to occur in advance of the leading edge of the phase-cut input voltage.

In a further embodiment of the present invention, an apparatus includes a load, a switching power converter, and a controller coupled to the switching power converter and the load. The controller is configured to predict a time period during a cycle of a phase-cut input voltage to a power converter system that is expected to occur in advance of a leading edge of the phase-cut input voltage. The controller is further configured to, during the cycle of the phase-cut input voltage, actively control a decreasing transition rate of a current conducted through a dimmer at least by the predicted time period that is expected to occur in advance of the leading edge of the phase-cut input voltage.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention may be better understood, and its numerous objects, features and advantages made apparent to those skilled in the art by referencing the accompanying drawings. The use of the same reference number throughout the several figures designates a like or similar element.

FIG. 1 (labeled prior art) depicts a lighting system that includes a leading edge dimmer.

FIG. 2 (labeled prior art) depicts exemplary voltage graphs associated with the lighting system of FIG. 1.

FIG. 3 depicts an electronic system that includes a controller to control current in accordance with adaptive timing and responsive current control.

FIG. 4 depicts an electronic system representing one embodiment of the electronic system of FIG. 3.

FIG. 5 depicts an adaptive timing and responsive current control process.

FIGS. 6-8 depict exemplary waveforms generated by the electronic system of FIG. 4 and the process of FIG. 6.

FIG. 9 depicts an exemplary zero cross detector and leading edge (LE) timing module.

FIG. 10 depicts an exemplary responsive current control module.

DETAILED DESCRIPTION

In at least one embodiment, an electronic system adapts current control timing for half line cycle of a phase-cut input voltage and responsively controls a dimmer current in a power converter system. The adaptive current control time and responsive current control provides, for example, interfacing with a dimmer. The electronic system and method include a dimmer, a switching power converter, and a controller to control the switching power converter and controls a dimmer current. In at least one embodiment, the controller determines a predicted time period from a zero crossing until a leading edge of a phase-cut input voltage and then responsively controls the dimmer current to, for example, reduce current and voltage perturbations (referred to as “ringing”), improve efficiency, and reduce an average amount of power handled by various circuit components.

For a triac-based dimmer during a period (referred to as “T_(OFF)”) of a phase-cut input voltage half line cycle from the time the half line cycle reaches a zero crossing until reaching a leading edge of a phase-cut input voltage, the dimmer does not conduct and, thus, phase cuts the supply voltage prior to conducting. During the non-conduction period T_(OFF), to properly recharge timing circuitry of the dimmer, the dimmer current has a glue value and is sometimes referred to in this non-conduction phase as a “glue current”. The glue value varies by dimmer from, for example, 10 mA to 300 mA. When the output voltage of the dimmer (referred to as phase-cut voltage “V_(φ) _(—) _(DIM)”) reaches a firing voltage V_(F) level, the dimmer fires (i.e. begins conducting) and conducts a dimmer current having a firing value and is .sometimes referred to at this event as a “firing current.” A typical firing value is 5 mA-50 mA. Thus, the firing value is often less than the glue value. Conventionally, the dimmer current virtually instantly transitions from the glue current value to the firing current value. Accordingly, a steep dimmer current transition occurs when the dimmer current transitions from the glue current to the firing current. The steep transition can result in undesirable ringing in a resistor-inductor-capacitor (“RLC network”) in the electronic system.

In at least one embodiment, the controller monitors a phase-cut input voltage to determine when leading edges of the phase cut voltage occur. The controller utilizes the history of when one or more leading edges occurred to predict a time period during a current or future cycle of the phase-cut input voltage to a. switching power converter that is expected to occur in advance of a leading edge of the phase-cut input voltage Then, during a cycle of the phase-cut input voltage, the controller actively controls a decreasing transition rate of the dimmer current conducted through the dimmer at least by the predicted time period that is expected to occur in advance of the leading edge of the phase-cut input voltage.

FIG. 3 depicts an electronic system 300 that includes a controller 302 to generate a control signal CS to control a switching power converter 304 of a power converter system 306. The controller 302 controls current in accordance with adaptive timing and responsive current control. A voltage supply 301 supplies a supply voltage V_(SUPPLY) to the electronic system 300. In at least one embodiment, the voltage supply 301 can be any voltage supply sufficient to provide power to load 316 and, in at least one embodiment, is the same as voltage supply 104. The dimmer 312 phase cuts the supply voltage V_(SUPPLY) to generate the phase cut input voltage V_(φ) _(—) _(DIM) to the power converter system 306. In at least one embodiment, the dimmer 312 is a triac-based dimmer, such as dimmer 102. The switching power converter 304 can be any type of switching power converter such as a boost, flyback, boost-buck, or Cúk type switching power converter. The load 316 can be any type of load, such as a load that includes one or more light emitting diodes.

The controller 302 includes a predictive current control timing module 308 to predict a time period in advance of when a leading edge, such as leading edge 204 (FIG. 2) of a phase-cut input voltage V_(φ) _(—) _(DIM) to the power converter system 306, or to a rectified version of the phase-cut input voltage V_(φ) _(—) _(DIM), is expected to occur. The controller 302 also includes a responsive current control module 310 that receives a leading edge prediction signal LEP. The responsive current control module 310 utilizes the leading edge prediction signal LEP to actively control a decreasing transition rate of the dimmer current i_(DIM) conducted through the dimmer 312 at least by the predicted time period that is expected to occur in advance of the leading edge of the phase-cut input voltage V_(φ) _(—) _(DIM). As subsequently described in more detail, by actively controlling the transition rate of the dimmer current i_(DIM), in at least one embodiment, potential ringing in the resistor-inductor-capacitor (“RLC”) network 314 is reduced.

FIG. 4 depicts an electronic system 400, which represents one embodiment of the electronic system 300. The power converter system 401, which represents one embodiment of the power converter system 306, includes a full bridge rectifier 402 that rectifies the phase-cut input voltage V_(φ) _(—) _(DIM) to generate the rectified, phase-cut input voltage V_(φ) _(—) _(R). Parasitic resistor 404, inductor 406, and capacitors 408 and 410 represent an RLC network 412, which represents one embodiment of RLC network 314. The switching power converter 304 receives a link voltage V_(L) and generates an output voltage V_(OUT) for load 316.

The electronic system 400 includes a. controller 414, which includes a predictive current control timing module 416. Predictive current control timing module 416 represents one embodiment of predictive current control timing module 308 (FIG. 3). The predictive current control timing module 416 predicts a time period in advance of when a leading edge, such as leading edge 204 (FIG. 2) of the phase-cut input voltage V_(φ) _(—) _(R) is expected to occur. The controller 414 also includes a responsive current control module 418 that receives a leading edge prediction signal LEP. The responsive current control module 418 utilizes the leading edge prediction signal LEP to actively control a decreasing transition rate of the dimmer current i_(DIM) conducted through the dimmer 312 at least by the predicted time period that is expected to occur in advance of the leading edge of the phase-cut input voltage V_(φ) _(—) _(R).

FIG. 5 depicts an adaptive timing and responsive current control process 500. Referring to FIGS. 4 and 5, in at least one embodiment, the predictive current control timing module 416 and responsive current control module 418 operate in accordance with the adaptive timing and responsive current control process 500. FIG. 6 depicts exemplary waveforms 600 for two half-line cycles HLC(n−1) and HLC(n) of the phase-cut input voltage V_(φ) _(—) _(R) and rectified dimmer current i_(φ) _(—) _(R). “n” is an index reference, and HLC(n) is a current half line cycle of the phase-cut input voltage V_(φ) _(—) _(R), HLC(n−1) is an immediately preceding half-line cycle of the phase-cut input voltage V_(φ) _(—) _(R), and so on. Referring to FIGS. 4, 5. and 6, in operation 502, the zero cross detector and leading edge (LE) timing module 420 detects zero crossings for (y-x) half line cycles HLC(n-x) through HLC(n-y), where “y” is a positive integer greater than or equal to “x”, which is also a positive integer. A “zero crossing” is when the phase-cut input voltage V_(φ) _(—) _(RV) _(R) is approximately 0V. The LE predictor 422 utilizes the elapsed time between the zero crossings and leading edges of half-line cycles to predict when the leading edge of the current half line cycle HLC(n) will occur. The particular values of “x” and “y” are a matter of design choice.

In at least one embodiment, and as depicted in FIG. 6, the values of “x” and “y” are both 1, which indicates that the zero cross detector and LE timing module 420 utilizes the immediately preceding half line cycle HLC(n−1) to predict the occurrence of the leading edge of the current half-line cycle HLC(n). In at least one embodiment, the zero cross detector and leading edge (LE) timing module 420 detects the zero crossing ZC(n−1) of the half line cycle HLC(n−1). In operation 504, the responsive current control module 418 sets the dimmer current i_(φ) _(—) _(R) to a predetermined “glue value” upon detection of the zero crossing ZC(n−1). When the dimmer current i_(φ) _(—) _(R) is set to the glue value, the dimmer current i_(φ) _(—) _(R) is sometimes referred to as a “glue current”. Exemplary glue values are 10 mA to 300 mA. In operation 506, the zero cross detector and leading edge (LE) timing module 420 detects the leading edge time for half line cycles HLC(n-x) through HLC(n-y). In at least one embodiment, operation 506 determines the occurrence of the leading edge LE(n−1) for HLC(n−1). Upon detection of the zero crossing ZC(n−1) and the occurrence of the leading edge LE(n−1) for the half line cycle HLC(n−1), in operation 508, the zero cross detector and leading edge (LE) timing module 420 determines a time period T_(OFF(n))−1 representing the elapsed time between the occurrences of ZC(n−1) and LE(n−1). The zero cross detector and leading edge (LE) timing module 420 provides the time period T_(OFF(n-1)) to the LE predictor 422. In operation 508, LE predictor 422 predicts the leading edge occurrence time period of the half line cycle HLC(n) as T_(OFF(n-1)) minus an offset value T_(OS(n)), and the subtraction result is referred to as the “predicted T_(OFF(n)P)”. In at least one embodiment, the LE predictor 422 stores the predicted T_(OFF(n)P) in a memory (not shown), and the responsive current control module 418 retrieves the predicted T_(OFF(n)) for the memory. In at least one embodiment, the LE predictor 422 provides the predicted T_(OFF(n)P) to the responsive current control module 422.

The offset value T_(OS(n)) provides a margin of error for a leading edge in the current half line cycle HLC(n) that occurs earlier than a leading edge of the immediately preceding leading edge of the line cycle HLC(n−1) relative to the zero crossings of the half line cycles. The particular choice of the offset value T_(OS(n)) is a matter of design choice. In at least one embodiment, the offset value T_(OS(n)) is set to a fixed value such as 400 μsec. In at least one embodiment, the LE predictor 422 determines the offset value T_(OS(n)) based on the measured dimmer non-conductive durations for N previous half line cycles, where N is an integer greater than or equal to 2. In at least one embodiment, the LE predictor 422 determines a trend of the actual durations of the dimmer non-conductive states and utilizes the trend to determine an offset value T_(OS(n)). In at least one embodiment, to utilize the trend to determine an offset value T_(OS(n)), the LE predictor 422 determines a rate of change of the actual durations of the dimmer non-conductive state time periods T_(OFF(n-1)A) through T_(OFF(n-N)A) for a set of N cycles of the phase-cut input voltage V_(φ) _(—) _(R) (or V₁₀₀ _(—) _(DIM)), and applies the rate of change to a previous offset T_(OS(n))−1 to determine the value of the current offset T_(OS(n)) so that the rate of change between the current offset T_(OS(n)) and the previous offset T_(OS(n-1)) is approximately the same as the change of the actual durations of the dimmer non-conductive states. In operation 510, the zero cross detector and leading edge (LE) timing module 420 determines the actual time period T_(OFF(n)A) and stores the value of the actual time period T_(OFF(n)A) in a memory (not shown) or provides the actual time period T_(OFF(n)A) to the responsive current control module 418.

The leading edge of the current half line cycle HLC(n) either occurs earlier, at the same time, or later than the leading edge of the immediately preceding half line cycle HLC(n−1) relative to the preceding zero crossing. In at least one embodiment, the leading edge occurs is a function of a dimmer setting of dimmer 312. As subsequently described in more detail, the responsive current control module 418 responds differently depending on whether the actual elapsed time period T_(OFF(n)A) between the zero crossing ZC(n) and the leading edge LE(n) of the current half line cycle HLC(n) is equal to or shorter than or longer than the predicted T_(OFF(n)P).

In operation 512, the responsive current control module 418 determines whether the actual time period T_(OFF(n)A) is greater than the predicted time period T_(OFF(n)P). If the actual time period T_(OFF(n)A) is greater than the predicted time period T_(OFF(n)P), the responsive current control module 418 performs operation 514 to transition the dimmer current i_(φ) _(—) _(R) from the glue value to a lower, firing current value. The tiring current value of the dimmer current i_(φ) _(—) _(R) is the value of the dimmer current i_(φ) _(—) _(R) when the dimmer 312 begins to conduct. The adaptive timing and responsive current control process 500 then returns to operation 502 and repeats for the next half line cycle of the phase-cut input voltage V_(φ) _(—) _(R).

Waveforms 600 depict the operation 512 scenario of the actual elapsed time period T_(OFF(n)A) between the zero crossing ZC(n) and the leading edge LE(n) of the current half line cycle HLC(n) is longer than the predicted elapsed time T_(OFF(n)P) between the zero crossing ZC(n) and the leading edge LE(n) of the current half line cycle HLC(n) minus an offset T_(OS(n)). In operation 512, the responsive current control module 418 actively controls the dimmer current i_(φ) _(—) _(R) to transition the dimmer current i_(φ) _(—) _(R) from the glue value to the firing value. The particular rate of transition is a matter of design choice. In at least one embodiment, the rate of transition is fast enough that dimmer current i_(φ) _(—) _(R) should reach the firing value prior to the actual occurrence of the leading edge of the half line cycle HLC(n). In at least one embodiment, the firing value equals an “attach current” value and is, for example, 50 mA. An attach state begins at the leading edge LE(n) and occurs during an initial charge transfer period from the leading edge LE(n). In at least one embodiment, without the attach current, if the dimmer current i_(φ) _(—) _(R) transitions all the way to the holding current value prior to the leading edge LE(n), the power converter system 402 can present a large enough input impedance that inadequately damps the RLC network 412.

FIG. 7 depicts exemplary waveforms 700 for the current half cycle HLC(n) when, if superimposed, the current leading edge LE(n) would occur prior to the previous leading edge LE(n−1). Referring to FIGS. 4, 5, and 7, in operation 512, if the zero cross detector and leading edge (LE) timing module 420 detects the current leading edge LE(n) and the responsive current control module 418 determines the actual time period T_(OFF(n)A) is less than or equal to the predicted T_(OFF(n)P) as depicted in FIG. 7, the dimmer 312 began conducting prior to an end of the predicted time period T_(OFF(n)P). Thus, the glue value for dimmer current i_(φ) _(—) _(R) is the same as the firing value as indicated in operation 516. In operation 518, the responsive current control module 418 transitions the dimmer current i_(φ) _(—) _(R) to the attach current value and then to the holding current value. The adaptive timing and responsive current control process 500 then returns to operation 502 and repeats for the next half line cycle of the phase-cut input voltage i_(φ) _(—) _(R).

FIG. 8 depicts exemplary waveforms 800 for the current half cycle HLC(n) when, if superimposed, the current leading edge LE(n) would occur prior to the previous leading edge LE(n−1). Referring to FIGS. 4, 5, and 8, the phase-cut input voltage V_(φ) _(—) _(R) is identical for waveforms 700 and 800. However, when in operation 512, the zero cross detector and leading edge (LE) timing module 420 detects the current leading edge LE(n) and the responsive current control module 418 determines the actual time period T_(OFF(n)A) is less than or equal to the predicted T_(OFF(n)P), in operation 518, and the responsive current control module 418 transitions the dimmer current i_(φ) _(—) _(R) to a higher attach current value than the attach current value in the dimmer current i_(φ) _(—) _(R) in waveforms 700. The higher attach current value provides a smoother transition from the glue value to the holding value for the dimmer current i_(φ) _(—) _(R). In at least one embodiment, the smoother transition assists in preventing a premature disconnect of the dimmer 312. The adaptive timing and responsive current control process 500 then returns to operation 502 and repeats for the next half line cycle of the phase-cut input voltage V_(φ) _(—) _(R). Additionally, in at least one embodiment, not only does the attach current value increase, in at least one embodiment, the responsive current control module 418 actively controls the duration of the dimmer current i_(φ) _(—) _(R) at the attach current, and generally increases the duration, to help ensure a smooth transition for the dimmer current i_(φ) _(—) _(R) from the attach value to the holding value.

FIG. 9 depicts zero cross detector and leading edge (LE) timing module 900, which represents one embodiment of zero cross detector and leading edge (LE) timing module 420. Comparator 902 compares a sensed version of the phase-cut input voltage V_(φ) _(—) _(R) to a known threshold value V_(TH). In at least one embodiment, the threshold value within the range of 0-10V, such as 5V. When the phase-cut input voltage V_(φ) _(—) _(R) is greater than the threshold value, the output ZC:LE of the comparator 902 is a logical 0. When the phase-cut input voltage falls below the threshold value V_(TH), which indicates a zero crossing, the output ZC:LE transitions from a logical 0 to a logical 1. When the output ZC:LE transitions from a logical 0 to a logical 1, the timer 904 begins counting pulses of a periodic clock signal f_(CLK) having a known frequency to timer 904. When the dimmer 312 transitions from nonconductive to conductive, a leading edge of the phase-cut input voltage V_(φ) _(—) _(R) occurs, and the phase-cut input voltage V_(φ) _(—) _(R) rises. When the phase-cut input voltage rises above the threshold value V_(TH), the output ZC:LE of the comparator 902 transitions from a logical 1 to a logical 0. At the transition from a logical 1 to a logical 0, the timer 904 marks the number of elapsed clock signal f_(CLK) pulses since the previous zero crossing detection. The elapsed number of clock signal f_(CLK) pulses represent the actual time period T_(OFF(n)A). The timer 904 then resets to determine the actual time period for the next half line cycle of the phase-cut input voltage V_(φ) _(—) _(R).

FIG. 10 depicts a responsive current control module 1000, which represents one embodiment of the responsive current control module 418. The current control logic 1002 receives the leading edge prediction signal LEP from the LE predictor 422 and performs operations 512 and 516. To control the dimmer current i_(φ) _(—) _(R) transitions in operations 514 and 518, the control logic 1002 controls a digital current control value iDIM_CONTROL. The current control value iDIM_CONTROL is an M+1 bit signal having bits [B₀. B₁, . . . , B_(M)], and M is a positive integer, such as 4, 8, or 16. The digital current control value iDIM_CONTROL is an input to a current source 1001, which controls the value of the dimmer current i_(φ) _(—) _(R).

During operation, current source 1001 sources current from source voltage node 407 and provides a variable impedance path for dimmer current i_(φ) _(—) _(R) to control the value of the dimmer current i_(φ) _(—) _(R). Current source 1001 includes a bias current source 1003 that generates a bias current i_(BIAS). A drain and gate of FET 1004 are connected together to form a “diode connected” configuration. The M+1 series connected FET pairs 1005.0/1006.0 through 1005.N/1006.N are respectively configured in a current mirror arrangement with FET 1004 to mirror the bias current i_(BIAS). “M” is an integer, and the value of M is a matter of design choice. Each pair of FETs 1005.X/1006.X is sized so that each subsequent pair sources twice as much current as the previous pair, e.g. FET pair 1005.1/1006.1 sources twice as much current as FET pair 1005.0/1006.0, and so on. “X” is an integer index ranging from 0 to M. In at least one embodiment, the value of M determines a maximum level of current capable of being sourced through current source 1001.

In at least one embodiment, the variable impedance control signal I_VAR is a digital value having M+1 bits, i.e. I_VAR=[B₀, B₁, . . . , B_(M)]. Each bit B₀, B₁, . . . , B_(M) is applied to the gate of a respective FET pair 1005.0/1006.0, 1005.1/1006.1, . . . , 1005.M/1006.M to control conductivity of the FET pairs. To operate the current source 1001, boost controller CCM/CRM controller 1202 (FIG. 12) sets a. logical value of I_VAR to set bits [B₀, B₁,k . . . , B_(M)]. For example, to turn all of the FET pairs ON, boost controller CCM/CRM controller 1202 sets [B₀, B₁, . . . , B_(M)]=[1, 1, . . . , 1] to cause each FET pair 1005.0/1006.0, 1005.111006.1, . . . , 1005.M/1006.M to conduct and sets bits to a logical value of I_VAR to B₀, B₁, . . . , B_(M)=[0, 0, . . . , 0] to cause each FET pair 1005.0/1006.0, 1005.1/1006.1, . . . , 1005.M/1006.M to turn “off”, i.e. nonconductive. In at least one embodiment, to current control logic 1002 decreases the value of bits [B₀, B₁, . . . , B_(M)] so that the current i_(φ) _(—) _(R) follows the decreasing transition set as, for example, shown in FIGS. 6, 7, and 8.

Thus, in at least one embodiment, an electronic system adapts current control timing for half line cycle of a phase-cut input voltage and responsively controls a dimmer current in a power converter system. The adaptive current control time and responsive current control provides, for example, interfacing with a dimmer.

Although embodiments have been described in detail, it should be understood that various changes, substitutions, and alterations can be made hereto without departing from the spirit and scope of the invention as defined by the appended claims. 

1. A method comprising: predicting a time period during a cycle of a phase-cut input voltage to a power converter system that is expected to occur in advance of a leading edge of the phase-cut input voltage; and during the cycle of the phase-cut input voltage, actively controlling a decreasing transition rate of a current conducted through a dimmer at least by the predicted time period that is expected to occur in advance of the leading edge of the phase-cut input voltage. 2-28. (canceled) 